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? semiconductor components industries, llc, 2006 july, 2006 ? rev. 8 1 publication order number: ncp1442/d ncp1442, ncp1443, ncp1444, ncp1445 4.0 a 280 khz/560 khz boost regulators the ncp1442/3/4/5 products are 280 khz/560 khz switching regulators with a high efficiency, 4.0 a integrated switch. these parts operate over a wide input voltage range, from 2.7 v to 30 v. the flexibility of the design allows the chips to operate in most power supply configurations, including boost, flyback, forward, inverting, and sepic. the ics utilize current mode architecture, which allows excellent load and line regulation, as well as a practical means for limiting current. combining high ? frequency operation with a highly integrated regulator circuit results in an extremely compact power supply solution. the circuit design includes provisions for features such as frequency synchronization, shutdown, and feedback controls for either positive or negative voltage regulation. part number frequency feedback voltage polarity ncp1442 280 khz positive ncp1443 280 khz negative ncp1444 560 khz positive ncp1445 560 khz negative features ? pb ? free packages are available* ? integrated power switch: 4.0 a guaranteed ? wide input range: 2.7 v to 30 v ? high frequency allows for small components ? minimum external components ? easy external synchronization ? built ? in overcurrent protection ? frequency foldback reduces component stress during an overcurrent condition ? thermal shutdown with hysteresis ? regulates either positive or negative output voltages ? shut down current: 50 a maximum applications ? boost converter ? inverting converter ? distributed power ? portable computers ? battery powered systems *for additional information on our pb ? free strategy and soldering details, please download the on semiconductor soldering and mounting techniques reference manual, solderrm/d. ordering information see detailed ordering and shipping information in the package dimensions section on page 18 of this data sheet. powerflex 7 ? pin f suffix case 936j 1 7 ncp1442/4 pin 1. v c 2. fb 3. test 4. gnd 5. v sw 6. ss 7. v cc ncp1443/5 1. v c 2. test 3. nfb 4. gnd 5. v sw 6. ss 7. v cc pin connections and marking diagrams x = device number 2, 3, 4, or 5 a = assembly location wl = wafer lot y = year ww = work week nc p144xf awlyww nc p144xt awlyww 1 7 17 17 7 lead, to ? 220 t suffix case 821p powerflex 7 ? pin 7 lead, to ? 220 http://onsemi.com
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 2 v sw gnd ss 3.3 v + ncp1442/4 1 2 3 4 5 6 7 5 v out /1.5 a 33 f v c nc ss gnd fb v sw + mbrs320t3 220 pf 22 k 5.1 k 0.01 f v cc 33 f 33 f 7.5 k gnd 10 h + figure 1. application diagram ? ncp1442/4, 3.3 v to 5.0 v/1.5 a boost converter 33 f + 33 f + + 33 f maximum ratings rating value unit thermal resistance junction ? to ? air, to220 ? 7 version in air (socketed) thermal resistance junction ? to ? air, to220 ? 7 version on cold plate (25 c) 66.7 1.45 c/w thermal resistance junction ? to ? air, powerflex on 2.1 sq. in. 1 oz. 53.8 c/w junction temperature range, t j 0 to +150 c storage temperature range, t storage ? 65 to +150 c lead temperature soldering: reflow (note 1) 230 peak c esd, human body model 2.0 kv stresses exceeding maximum ratings may damage the device. maximum ratings are stress ratings only. functional operation above t he recommended operating conditions is not implied. extended exposure to stresses above the recommended operating conditions may af fect device reliability. 1. 60 second maximum above 183 c. maximum ratings pin name pin symbol v max v min i source i sink ic power input v cc 30 v ? 0.3 v n/a 200 ma shutdown/sync ss 30 v ? 0.3 v 1.0 ma 1.0 ma loop compensation v c 6.0 v ? 0.3 v 10 ma 10 ma voltage feedback input fb (ncp1442/4 only) 10 v ? 0.3 v 1.0 ma 1.0 ma negative feedback input (transient, 10 ms) nfb (ncp1443/5 only) 10 v ? 10 v 1.0 ma 1.0 ma test pin test 6.0 v ? 0.3 v 1.0 ma 1.0 ma ground gnd 0.3 v ? 0.3 v 9.0 a 10 ma switch input v sw 40 v ? 0.3 v 10 ma 9.0 a
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 3 electrical characteristics (2.7 v < v cc < 30 v; 0 c < t a < 85 c; 0 c < t j < 125 c; for all ncp1442/3/4/5 specifications unless otherwise stated.) (see note 2) characteristic test conditions min typ max unit positive and negative error amplifiers fb reference voltage (ncp1442/4 only) v c tied to fb; measure at fb 1.246 1.276 1.300 v nfb reference voltage (ncp1443/5 only) v c = 1.25 v ? 2.60 ? 2.475 ? 2.40 v fb input current (ncp1442/4 only) fb = v ref ? 1.0 0.1 1.0 a nfb input current (ncp1443/5 only) nfb = nv ref ? 16 ? 10 ? 5.0 a fb reference voltage line regulation (ncp1442/4 only) v c = fb ? 0.03 0.01 0.03 %/v nfb reference voltage line regulation (ncp1443/5 only) v c = 1.25 v ? 0.05 0.01 0.05 %/v positive error amp transconductance i vc = 25 a 300 550 800 mho negative error amp transconductance i vc = 5.0 a 115 160 225 mho positive error amp gain (note 3) 200 500 ? v/v negative error amp gain (note 3) 100 180 320 v/v v c source current fb = 1.0 v or nfb = ? 1.9 v, v c = 1.25 v ? 90 ? 50 ? 25 a v c sink current fb = 1.5 v or nfb = ? 3.1 v, v c = 1.25 v 200 460 1500 a v c high clamp voltage fb = 1.0 v or nfb = ? 1.9 v; v c sources 25 a 1.5 1.64 1.9 v v c low clamp voltage fb = 1.5 v or nfb = ? 3.1 v, v c sinks 25 a 0.30 0.47 0.70 v v c threshold reduce v c from 1.5 v until switching stops 0.70 1.05 1.30 v oscillator base operating frequency ncp1442/3, fb = 1.0 v or nfb = ? 1.9 v 240 280 320 khz reduced operating frequency ncp1442/3, fb = 0 v or nfb = 0 v 30 68 120 khz maximum duty cycle ncp1442/3 90 96 ? % base operating frequency ncp1444/5, fb = 1.0 v or nfb = ? 1.9 v 480 560 640 khz reduced operating frequency ncp1444/5, fb = 0 v or nfb = 0 v 60 120 160 khz maximum duty cycle ncp1444/5 82 92 ? % fb frequency shift threshold frequency drops to reduced operating frequency 0.36 0.40 0.44 v nfb frequency shift threshold frequency drops to reduced operating frequency ? 0.80 ? 0.68 ? 0.50 v sync/shutdown sync range ncp1442/3 ? 500 ? khz sync range ncp1444/5 ? 1000 ? khz sync pulse transition threshold rise time = 20 ns ? 2.5 ? v ss bias current ss = 0 v ss = 3.0 v ? 10 ? ? 1.0 0.2 ? 4.0 a a shutdown threshold ? 0.50 0.85 1.20 v shutdown delay 2.7 v v cc 12 v 12 v < v cc 30 v 12 12 100 40 500 400 s s 2. for the fr4 suffix parts, production testing is performed at 25 c and 85 c; limits at 0 c are guaranteed by design. 3. guaranteed by design, not 100% tested in production.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 4 electrical characteristics (continued) (2.7 v < v cc < 30 v; 0 c < t a < 85 c; 0 c < t j < 125 c; for all ncp1442/3/4/5 specifications unless otherwise stated.) (see note 2) characteristic test conditions min typ max unit power switch switch saturation voltage i switch = 4.0 a i switch = 10 ma, 2.7 v < v cc < 12 v i switch = 10 ma, 12 v < v cc < 30 v ? ? ? 0.6 0.14 0.9 1.0 0.5 0.4 v v v switch current limit 50% duty cycle (note 4) 80% duty cycle (note 4) 5.0 4.0 6.0 ? 8.0 ? a a minimum pulse width fb = 0 v or nfb = 0 v, i sw = 4.0 a (note 4) 200 250 300 ns switch transconductance, i cc / iv sw 2.7 v v cc 12 v, 10 ma i sw 4.0 a 12 v < v cc 30 v, 10 ma i sw 4.0 a ? ? 8.0 10 30 50 ma/a switch leakage v sw = 40 v, v cc = 0v ? 2.0 20 a general operating current i sw = 0 ? 15 27 ma shutdown mode current v c < 0.8 v, ss = 0 v, 2.7 v v cc 12 v v c < 0.8 v, ss = 0 v, 12 v v cc 30 v ? ? 16 25 60 60 a minimum operation input voltage v sw switching, maximum i sw = 10 ma ? 2.2 2.6 v thermal shutdown (note 4) 150 180 210 c thermal hysteresis (note 4) ? 25 ? c 4. guaranteed by design, not 100% tested in production. package pin description package pin number pin symbol function 1 v c loop compensation pin. the v c pin is the output of the error amplifier and is used for loop compensation, current limit and soft start. loop compensation can be implemented by a sim- ple rc network as shown in the application diagram on page 2. 2 (ncp1442/4 only) fb positive regulator feedback pin. this pin senses a positive output voltage and is referenced to 1.276 v. when the voltage at this pin falls below 0.4 v, chip switching frequency reduces to 20% of the nominal frequency. 2 (ncp1443/5 only) 3 (ncp1442/4 only) test these pins are connected to internal test logic and should either be left floating or tied to ground. connection to a voltage between 2.0 v and 6.0 v shuts down the internal oscillator and leaves the power switch running. 3 (ncp1443/5 only) nfb negative feedback pin. this pin senses a negative output voltage and is referenced to ? 2.475 v. when the voltage at this pin goes above ? 0.65 v, chip switching frequency reduces to 20% of the nominal frequency. 4 gnd ground pin. this pin provides a ground for the controller circuitry and the internal power switch. this pin is internally connected to the metal pad of the package to provide an addition- al ground connection as well as an effective means of dissipating heat. 5 v sw high current switch pin. this pin connects internally to the collector of the power switch. the open voltage across the power switch can be as high as 40 v. to minimize radiation, use a trace as short as practical. 6 ss synchronization and shutdown pin. this pin may be used to synchronize the part to nearly twice the base frequency. a ttl low will shut the part down and put it into low current mode. if synchronization is not used, this pin should be either tied high or left floating for normal operation. 7 v cc input power supply pin. this pin supplies power to the part and should have a bypass capaci- tor connected to gnd.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 5 gnd v sw + ? + ? + ? v cc ss nfb fb positive error amp ncp1443/5 only ncp1442/4 only negative error amp pwm comparator ramp summer slope compensation thermal shutdown 2.0 v regulator delay timer sync shutdown oscillator frequency shift 5:1 s pwm latch r q driver switch 15 m ? 0.65 v detector 0.4 v detector 1.276 v 250 k 200 k 2.0 v v c 5 figure 2. block diagram
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 6 t a , ambient temperature ( c) 0204060 265 t a , ambient temperature ( c) 40 60 20 080 255 270 250 260 275 290 10 15 0 5 20 figure 3. supply current versus temperature v cc = 30 v figure 4. i cc / i sw versus temperature figure 5. switch saturation voltage versus switch current figure 6. switching frequency versus temperature (ncp1442/3 only) 12 8 6 10 0 t a , ambient temperature ( c) i cc / i sw (ma/a) 40 60 20 080 4 400 0 1.0 0.5 100 500 0 200 600 800 700 300 1.5 2.0 2.5 3.0 3.5 4.0 i sw , switch current (a) v ce(sat) , switch saturation voltage (mv) 2 i cc , supply current (ma) f, switching frequency (khz) 80 v fb , positive feedback voltage (v) 0.42 0.43 0.41 0.40 0.45 0.39 0 25 100 125 figure 7. switching frequency versus temperature (ncp1444/5 only) figure 8. switching frequency versus positive feedback voltage 555 585 595 590 t a , ambient temperature ( c) f, switching frequency (khz) 0.38 f, switching frequency (% of typical) 560 565 570 020 50 75 v cc = 12 v v cc = 2.7 v v cc = 30 v v cc = 12 v v cc = 2.7 v i sw = 2.99 a v cc = 2.7 v t a = 25 c t a = 85 c 280 285 40 60 80 575 580 v cc = 12 v t a = 25 c t a = 85 c 0.44 300 295 30 50 10 70 550 600 30 50 10 70
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 7 ? 0.665 ? 0.67 v nfb , negative feedback voltage (v) ? 0.675 ? 0.68 ? 0.685 ? 0.69 0.23 t a , ambient temperature ( c) 40 60 20 080 0.21 0.25 0.20 0.22 0.26 0.30 50 75 0 25 125 figure 9. switching frequency versus negative feedback voltage figure 10. feedback reference voltage versus temperature (ncp1442/4 only) figure 11. feedback reference voltage versus temperature (ncp1443/5 only) figure 12. error amplifier bias current versus temperature (ncp1442/3 only) 1.269 t a , ambient temperature ( c) v fb , feedback reference voltage (v) 40 60 20 080 ? 2.47 ? 2.50 ? 2.49 ? 2.46 020406080 t a , ambient temperature ( c) v fb , feedback reference voltage (v) 1.270 f, switching frequency (% of typical) i fb , error amplifier bias current ( a) ? 0.695 t a , ambient temperature ( c) 20 40 0 80 93.0 93.5 97.0 figure 13. error amplifier bias current versus temperature (ncp1443/5 only) figure 14. maximum duty cycle versus temperature ? 14 ? 10 ? 8 ? 9 t a , ambient temperature ( c) i nfb , error amplifier bias current ( a) d max , maximum duty cycle (%) ? 13 ? 12 020 94.0 95.0 v cc = 30 v v cc = 12 v v cc = 2.7 v v cc = 2.7 v 0.28 0.29 40 60 80 ? 11 v cc = 2.7 v 60 100 v cc = 12 v t a = 25 c t a = 85 c 1.271 1.272 1.273 1.274 1.275 1.276 v cc = 30 v v cc = 12 v ? 2.48 0.24 0.27 v cc = 2.7 v v cc = 30 v v cc = 12 v v cc = 30 v 12 v 2.7 v 94.5 95.5 96.0 96.5 v cc = 12 v v cc = 30 v 30 50 10 70 1.268
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 8 t a , ambient temperature ( c) 0204060 t a , ambient temperature ( c) 40 60 20 080 0 50 150 1.08 1.02 1.04 1.00 figure 15. v c threshold voltage versus temperature figure 16. shutdown threshold versus temperature figure 17. shutdown delay versus temperature (ncp1442) figure 18. shutdown delay versus temperature (ncp1444) 0 t a , ambient temperature ( c) v ss , shutdown threshold (v) 40 60 20 080 60 0 20 100 160 120 020406080 t a , ambient temperature ( c) t d , shutdown delay ( s) 0.3 v cth , threshold voltage (v) t d , shutdown delay ( s) 80 v cc , supply voltage (v) 10 20 15 30 5 0 5 45 figure 19. i ss versus v ss figure 20. supply current versus supply voltage during shutdown 0 05 3.0 4.0 3.5 v ss (v) i ss ( a) 0 i sd , supply current ( a) 0.5 1.0 10 15 10 20 v cc = 2.7 v 200 250 20 25 30 2.5 25 1.14 0.4 0.6 0.8 1.0 0.9 v cc = 30 v v cc = 12 v 40 100 v cc = 2.7 v v cc = 30 v v cc = 12 v 15 25 35 40 1.06 1.10 1.12 80 140 2.0 1.5 30 t a = 25 c t a = 85 c 30 50 10 70 30 50 10 70 30 50 10 70 0.7 0.5 0.2 0.1 180 30 50 10 70
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 9 t a , ambient temperature ( c) 0204060 v ref ? v nfb , feedback voltage (mv) 0 0.1 ? 0.1 0.2 ? 0.2 ? 200 ? 150 ? 50 100 530 470 480 570 figure 21. error amplifier transconductance versus temperature figure 22. negative error amplifier transconductance versus temperature figure 23. error amplifier output current versus positive feedback voltage figure 24. error amplifier output current versus negative feedback voltage ? 950 t a , ambient temperature ( c) gm, transconductance ( mho) 40 60 20 080 ? 300 ? 0.2 ? 0.25 100 ? 100 ? 0.15 0 0.05 0.2 0.25 v ref ? v fb , feedback voltage (mv) i c , ea output current ( a) ? 900 gm, transconductance ( mho) ? 0.3 i c , ea output current ( a) 80 figure 25. switch leakage current versus temperature 3.0 5.0 6.0 5.5 t a , ambient temperature ( c) i sw , switch leakage current ( a) 3.5 020 0 50 40 60 80 4.5 560 ? 850 ? 800 ? 700 ? 600 ? 500 ? 100 490 540 550 ? 200 0 4.0 500 510 520 ? 650 ? 750 0.1 0.15 ? 0.1 ? 0.05 ? 400
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 10 applications information theory of operation current mode control + ? driver c o r loa d v sw x5 summer slope compensation v c oscillator d1 v cc s r q in out pwm comparator l 15 m figure 26. current mode control scheme power switch the ncp144x family incorporates a current mode control scheme, in which the pwm ramp signal is derived from the power switch current. this ramp signal is compared to the output of the error amplifier to control the on ? time of the power switch. the oscillator is used as a fixed ? frequency clock to ensure a constant operational frequency. the resulting control scheme features several advantages over conventional voltage mode control. first, derived directly from the inductor, the ramp signal responds immediately to line voltage changes. this eliminates the delay caused by the output filter and error amplifier, which is commonly found in voltage mode controllers. the second benefit comes from inherent pulse ? by ? pulse current limiting by merely clamping the peak switching current. finally, since current mode commands an output current rather than voltage, the filter offers only a single pole to the feedback loop. this allows both a simpler compensation and a higher gain ? bandwidth over a comparable voltage mode circuit. without discrediting its apparent merits, current mode control comes with its own peculiar problems, mainly, subharmonic oscillation at duty cycles over 50%. the ncp144x family solves this problem by adopting a slope compensation scheme in which a fixed ramp generated by the oscillator is added to the current ramp. a proper slope rate is provided to improve circuit stability without sacrificing the advantages of current mode control. oscillator and shutdown figure 27. timing diagram of sync and shutdown v sw current ramp sync the oscillator is trimmed to guarantee frequency accuracy. the output of the oscillator turns on the power switch at a frequency of 280 khz (ncp1442/3) or 560 khz (ncp1444/5), as shown in figure 26. the power switch is turned off by the output of the pwm comparator. a ttl ? compatible sync input at the ss pin is capable of syncing up to 1.8 times the base oscillator frequency. as shown in figure 27, in order to sync to a higher frequency, a positive transition turns on the power switch before the output of the oscillator goes high, thereby resetting the oscillator. the sync operation allows multiple power supplies to operate at the same frequency. a sustained logic low at the ss pin will shut down the ic and reduce the supply current. an additional feature includes frequency shift to 20% of the nominal frequency when either the nfb or fb pins trigger the threshold. during power up, overload, or short circuit conditions, the minimum switch on ? time is limited by the pwm comparator minimum pulse width. extra switch off ? time reduces the minimum duty cycle to protect external components and the ic itself. as previously mentioned, this block also produces a ramp for the slope compensation to improve regulator stability. error amplifier + ? + ? ncp1443/5 ncp1442/4 figure 28. error amplifier equivalent circuit 2.0 v 200 k 250 k 1m positive error ? amp negative error ? amp 1.276 v fb nfb v c c1 r1 5 k 0.01 f voltage clamp 120 pf for ncp1443/5, the nfb pin is internally referenced to ? 2.475 v with approximately a 250 k input impedance. for ncp1442/4, the fb pin is directly connected to the inverting input of the positive error amplifier, whose non ? inverting input is fed by the 1.276 v reference. both amplifiers are transconductance amplifiers with a high output impedance of approximately 1.0 m , as shown in figure 28. the v c pin is connected to the output of the error amplifiers and is internally clamped between 0.5 v and 1.7 v. a typical connection at the v c pin includes a capacitor in series with a resistor to ground, forming a pole/zero for loop compensation. an external shunt can be connected between the v c pin and ground to reduce its clamp voltage. consequently, the current limit of the internal power transistor current is reduced from its nominal value.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 11 switch driver and power switch the switch driver receives a control signal from the logic section to drive the output power switch. the switch is grounded through emitter resistors (15 m total) to the gnd pin. the peak switching current is clamped by an internal circuit. the clamp current is guaranteed to be greater than 4.0 a and varies with duty cycle due to slope compensation. the power switch can withstand a maximum voltage of 40 v on the collector (v sw pin). the saturation voltage of the switch is typically less than 1.0 v to minimize power dissipation. short circuit condition when a short circuit condition happens in a boost circuit, the inductor current will increase during the whole switching cycle, causing excessive current to be drawn from the input power supply. since control ics don?t have the means to limit load current, an external current limit circuit (such as a fuse or relay) has to be implemented to protect the load, power supply and ics. in other topologies, the frequency shift built into the ic prevents damage to the chip and external components. this feature reduces the minimum duty cycle and allows the transformer secondary to absorb excess energy before the switch turns back on. figure 29. startup waveforms of circuit shown in the application diagram. load = 400 ma. i l v out v c v cc the ncp144x can be activated by either connecting the v cc pin to a voltage source or by enabling the ss pin. startup waveforms shown in figure 29 are measured in the boost converter demonstrated in the block diagram (figure 2). recorded after the input voltage is turned on, this waveform shows the various phases during the power up transition. when the v cc voltage is below the minimum supply voltage, the v sw pin is in high impedance. therefore, current conducts directly from the input power source to the output through the inductor and diode. once v cc reaches approximately 1.5 v, the internal power switch briefly turns on. this is a part of the ncp144x?s normal operation. the turn ? on of the power switch accounts for the initial current swing. when the v c pin voltage rises above the threshold, the internal power switch starts to switch and a voltage pulse can be seen at the v sw pin. detecting a low output voltage at the fb pin, the built ? in frequency shift feature reduces the switching frequency to a fraction of its nominal value, reducing the minimum duty cycle, which is otherwise limited by the minimum on ? time of the switch. the peak current during this phase is clamped by the internal current limit. when the fb pin voltage rises above 0.4 v, the frequency increases to its nominal value, and the peak current begins to decrease as the output approaches the regulation voltage. the overshoot of the output voltage is prevented by the active pull ? on, by which the sink current of the error amplifier is increased once an overvoltage condition is detected. the overvoltage condition is defined as when the fb pin voltage is 50 mv greater than the reference voltage. component selection frequency compensation the goal of frequency compensation is to achieve desirable transient response and dc regulation while ensuring the stability of the system. a typical compensation network, as shown in figure 30, provides a frequency response of two poles and one zero. this frequency response is further illustrated in the bode plot shown in figure 31. ncp1442/3/4/5 figure 30. a typical compensation network v c gnd c1 r1 c2 the high dc gain in figure 31 is desirable for achieving dc accuracy over line and load variations. the dc gain of a transconductance error amplifier can be calculated as follows: gain dc g m r o where: g m = error amplifier transconductance; r o = error amplifier output resistance 1.0 m .
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 12 the low frequency pole, f p1, is determined by the error amplifier output resistance and c1 as: f p1 1 2 c1r o the first zero generated by c1 and r1 is: f z1 1 2 c1r1 the phase lead provided by this zero ensures that the loop has at least a 45 phase margin at the crossover frequency. therefore, this zero should be placed close to the pole generated in the power stage which can be identified at frequency: f p 1 2 c o r load where: c o = equivalent output capacitance of the error amplifier 120pf; r load = load resistance. the high frequency pole, f p2 , can be placed at the output filter?s esr zero or at half the switching frequency. placing the pole at this frequency will cut down on switching noise. the frequency of this pole is determined by the value of c2 and r1: f p2 1 2 c2r1 one simple method to ensure adequate phase margin is to design the frequency response with a ? 20 db per decade slope, until unity ? gain crossover. the crossover frequency should be selected at the midpoint between f z1 and f p2 where the phase margin is maximized. figure 31. bode plot of the compensation network shown in figure 30 frequency (log) f p1 gain (db) dc gain f z1 f p2 negative voltage feedback since the negative error amplifier has finite input impedance as shown in figure 32, its induced error has to be considered. if a voltage divider is used to scale down the negative output voltage for the nfb pin, the equation for calculating output voltage is: v out 2.475 (r1 r2) r2 10 a r1 + ? figure 32. negative error amplifier and nfb pin 2 v 200 k negative error ? amp r p nfb r in ? v out r1 250 k r2 it is shown that if r1 is less than 10 k, the deviation from the design target will be less than 0.1 v. if the tolerances of the negative voltage reference and nfb pin input current are considered, the possible offset of the output v offset varies in the range of: 0.0.5 (r1 r2) r2 (15 a r1) v offset 0.0.5 (r1 r2) r2 (5 a r1) v sw voltage limit in the boost topology, v sw pin maximum voltage is set by the maximum output voltage plus the output diode forward voltage. the diode forward voltage is typically 0.5 v for schottky diodes and 0.8 v for ultrafast recovery diodes: v sw(max) v out(max) v f where: v f = output diode forward voltage. in the flyback topology, peak v sw voltage is governed by: v sw(max) v cc(max) (v out v f ) n where: n = transformer turns ratio, primary over secondary. when the power switch turns off, there exists a voltage spike superimposed on top of the steady ? state voltage. usually this voltage spike is caused by transformer leakage inductance charging stray capacitance between the v sw and gnd pins. to prevent the voltage at the v sw pin from exceeding the maximum rating, a transient voltage suppressor in series with a diode is paralleled with the primary windings. another method of clamping switch voltage is to connect a transient voltage suppressor between the v sw pin and ground.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 13 magnetic component selection when choosing a magnetic component, one must consider factors such as peak current, core and ferrite material, output voltage ripple, emi, temperature range, physical size and cost. in boost circuits, the average inductor current is the product of output current and voltage gain (v out /v cc ), assuming 100% energy transfer efficiency. in continuous conduction mode, inductor ripple current is: i ripple v cc (v out v cc ) (f)(l)(v out) where: f = 280 khz for ncp1442/3 and 560 khz for ncp1444/5. the peak inductor current is equal to average current plus half of the ripple current, which should not cause inductor saturation. the above equation can also be referenced when selecting the value of the inductor based on the tolerance of the ripple current in the circuits. small ripple current provides the benefits of small input capacitors and greater output current capability. a core geometry like a rod or barrel is prone to generating high magnetic field radiation, but is relatively cheap and small. other core geometries, such as toroids, provide a closed magnetic loop to prevent emi. input capacitor selection in boost circuits, the inductor becomes part of the input filter, as shown in figure 34. in continuous mode, the input current waveform is triangular and does not contain a large pulsed current, as shown in figure 33. this reduces the requirements imposed on the input capacitor selection. during continuous conduction mode, the peak to peak inductor ripple current is given in the previous section. as we can see from figure 33, the product of the inductor current ripple and the input capacitor?s effective series resistance (esr) determine the v cc ripple. in most applications, input capacitors in the range of 10 f to 100 f with an esr less than 0.3 work well up to a full 4.0 a switch current. v cc ripple figure 33. boost input voltage and current ripple waveforms i in i l + ? figure 34. boost circuit effective input filter v cc c in r esr i l i in the situation is different in a flyback circuit. the input current is discontinuous and a significant pulsed current is seen by the input capacitors. therefore, there are two requirements for capacitors in a flyback regulator: energy storage and filtering. to maintain a stable voltage supply to the chip, a storage capacitor larger than 20 f with low esr is required. to reduce the noise generated by the inductor, insert a 1.0 f ceramic capacitor between v cc and ground as close as possible to the chip. output capacitor selection figure 35. typical output voltage ripple v out ripple i l by examining the waveforms shown in figure 35, we can see that the output voltage ripple comes from two major sources, namely capacitor esr and the charging/discharging of the output capacitor. in boost circuits, when the power switch turns off, i l flows into the output capacitor causing an instant v = i in esr. at the same time, current i l ? i out charges the capacitor and increases the output voltage gradually. when the power switch is turned on, i l is shunted to ground and i out discharges the output capacitor. when the i l ripple is small enough, i l can be treated as a constant and is equal to input
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 14 current i in . summing up, the output voltage peak ? peak ripple can be calculated by: v out(ripple) (i in i out) (1 d) (c out) (f) i out d (c out )(f) i in esr the equation can be expressed more conveniently in terms of v cc , v out and i out for design purposes as follows: v out(ripple) i out (v out v cc ) (c out )(f) 1 (c out )(f) (i out )(v out )(esr) v cc the capacitor rms ripple current is: i ripple (i in i out ) 2 (1 d) (i out ) 2 (d) i out v out v cc v cc although the above equations apply only for boost circuits, similar equations can be derived for flyback circuits. reducing the current limit in some applications, the designer may prefer a lower limit on the switch current than 4.0 a. an external shunt can be connected between the v c pin and ground to reduce its clamp voltage. consequently, the current limit of the internal power transistor current is reduced from its nominal value. the voltage on the v c pin can be evaluated with the equation: v c i sw r e a v where: r e = .015 , the value of the internal emitter resistor; a v = 5.0 v/v, the gain of the current sense amplifier. since r e and a v cannot be changed by the end user, the only available method for limiting switch current below 4.0 a is to clamp the v c pin at a lower voltage. if the maximum switch or inductor current is substituted into the equation above, the desired clamp voltage will result. a simple diode clamp, as shown in figure 36, clamps the v c voltage to a diode drop above the voltage on resistor r3. unfortunately, such a simple circuit is not generally acceptable if v in is loosely regulated. figure 36. current limiting using a diode clamp v c d1 v cc r1 v in c2 c1 r2 r3 another solution to the current limiting problem is to externally measure the current through the switch using a sense resistor. such a circuit is illustrated in figure 37. ? + figure 37. current limiting using a current sense resistor v c r sense q1 v cc r1 v in c2 c1 r2 c3 output ground pgnd agnd the switch current is limited to: i switch(peak) v be(q1) r sense where: v be(q1) = the base ? emitter voltage drop of q1, typically 0.65 v.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 15 the improved circuit does not require a regulated voltage to operate properly. unfortunately, a price must be paid for this convenience in the overall efficiency of the circuit. the designer should note that the input and output grounds are no longer common. also, the addition of the current sense resistor, r sense , results in a considerable power loss which increases with the duty cycle. resistor r2 and capacitor c3 form a low ? pass filter to remove noise. subharmonic oscillation subharmonic oscillation (shm) is a problem found in current ? mode control systems, where instability results when duty cycle exceeds 50%. shm only occurs in switching regulators with a continuous inductor current. this instability is not harmful to the converter and usually does not affect the output voltage regulation. shm will increase the radiated em noise from the converter and can cause, under certain circumstances, the inductor to emit high ? frequency audible noise. shm is an easily remedied problem. the rising slope of the inductor current is supplemented with internal ?slope compensation? to prevent any duty cycle instability from carrying through to the next switching cycle. in the ncp144x family, slope compensation is added during the entire switch on ? time, typically in the amount of 180 ma/ s. in some cases, shm can rear its ugly head despite the presence of the onboard slope compensation. the simple cure to this problem is more slope compensation to avoid the unwanted oscillation. in that case, an external circuit, shown in figure 38, can be added to increase the amount of slope compensation used. this circuit requires only a few components and is ?tacked on? to the compensation network. figure 38. technique for increasing slope compensation v c r1 c2 c1 r2 r3 v sw c3 v sw the dashed box contains the normal compensation circuitry to limit the bandwidth of the error amplifier. resistors r2 and r3 form a voltage divider off of the v sw pin. in normal operation, v sw looks similar to a square wave, and is dependent on the converter topology. formulas for calculating v sw in the boost and flyback topologies are given in the section ?v sw voltage limit.? the voltage on v sw charges capacitor c3 when the switch is off, causing the voltage at the v c pin to shift upwards. when the switch turns on, c3 discharges through r3, producing a negative slope at the v c pin. this negative slope provides the slope compensation. the amount of slope compensation added by this circuit is i t v sw r 3 r 2 r 3 1 e (1 d) r 3 c 3 f sw f sw (1 d)r e a v where: i/ t = the amount of slope compensation added (a/s); v sw = the voltage at the switch node when the transistor is turned off (v); f sw = the switching frequency, typically 280 khz (ncp1442/3) or 560 khz (ncp1444/5) (hz); d = the duty cycle; r e = 0.015 , the value of the internal emitter resistor; a v = 5.0 v/v, the gain of the current sense amplifier. in selecting appropriate values for the slope compensation network, the designer is advised to choose a convenient capacitor, then select values for r2 and r3 such that the amount of slope compensation added is 100 ma/ s. then r2 may be increased or decreased as necessary. of course, the series combination of r2 and r3 should be large enough to avoid drawing excessive current from v sw . additionally, to ensure that the control loop stability is improved , the time constant formed by the additional components should be chosen such that: r 3 c 3
1 d f sw finally, it is worth mentioning that the added slope compensation is a trade ? off between duty cycle stability and transient response. the more slope compensation a designer adds, the slower the transient response will be, due to the external circuitry interfering with the proper operation of the error amplifier. soft ? start through the addition of an external circuit, a soft ? start function can be added to the ncp1442/3/4/5 family of components. soft ? start circuitry prevents the v c pin from slamming high during startup, thereby inhibiting the inductor current from rising at a high slope. this circuit, shown in figure 39, requires a minimum number of components and allows the soft ? start circuitry to activate any time the ss pin is used to restart the converter.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 16 figure 39. soft ? start v c r1 c2 c1 d2 d1 v cc c3 v in ss ss resistor r1 and capacitors c1 and c2 form the compensation network. at turn on, the voltage at the v c pin starts to come up, charging capacitor c3 through schottky diode d2, clamping the voltage at the v c pin such that switching begins when v c reaches the v c threshold, typically 1.05 v (refer to graphs for detail over temperature). v c v f(d2) v c3 therefore, c3 slows the startup of the circuit by limiting the voltage on the v c pin. the soft ? start time increases with the size of c3. diode d1 discharges c3 when ss is low. if the shutdown function is not used with this part, the cathode of d1 should be connected to v in . calculating junction temperature to ensure safe operation of the ncp1442/3/4/5, the designer must calculate the on ? chip power dissipation and determine its expected junction temperature. internal thermal protection circuitry will turn the part off once the junction temperature exceeds 180 c 30 . however, repeated operation at such high temperatures will ensure a reduced operating life. calculation of the junction temperature is an imprecise but simple task. first, the power losses must be quantified. there are three major sources of power loss on the ncp144x: ? biasing of internal control circuitry, p bias ? switch driver, p driver ? switch saturation, p sat the internal control circuitry, including the oscillator and linear regulator, requires a small amount of power even when the switch is turned off. the specifications section of this datasheet reveals that the typical operating current, i q , due to this circuitry is 5.5 ma. additional guidance can be found in the graph of operating current vs. temperature. this graph shows that iq is strongly dependent on input voltage, v in , and the ambient temperature, t a . then: p bias v in i q since the onboard switch is an npn transistor, the base drive current must be factored in as well. this current is drawn from the v in pin, in addition to the control circuitry current. the base drive current is listed in the specifications as i cc / i sw , or switch transconductance. as before, the designer will find additional guidance in the graphs. with that information, the designer can calculate: p driver v in i sw i cc i sw d where: i sw = the current through the switch; d = the duty cycle or percentage of switch on ? time. i sw and d are dependent on the type of converter. in a boost converter, i sw(avg) i load d i efficiency d v out v in v out in a flyback converter, i sw(avg) v out i load v in i efficiency d v out v out n s n p v in where: n s = number of turns in the transformer secondary winding. n p = number of turns in the transformer primary winding. the switch saturation voltage, v (ce)sat , is the last major source of on ? chip power loss. v (ce)sat is the collector ? emitter voltage of the internal npn transistor when it is driven into saturation by its base drive current. the value for v (ce)sat can be obtained from the specifications or from the graphs, as ?switch saturation voltage.? thus, p sat v (ce)sat i sw d finally, the total on ? chip power losses are: p d p bias p driver p sat
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 17 power dissipation in a semiconductor device results in the generation of heat in the junctions at the surface of the chip. this heat is transferred to the surface of the ic package, but a thermal gradient exists due to the resistive properties of the package molding compound. the magnitude of the thermal gradient is expressed in manufacturers? data sheets as ja , or junction ? to ? ambient thermal resistance. the on ? chip junction temperature can be calculated if ja , the air temperature near the surface of the ic, and the on ? chip power dissipation are known. t j t a (p d ja ) where: t j = ic or fet junction temperature ( c); t a = ambient temperature ( c); p d = power dissipated by part in question (w); ja = junction ? to ? ambient thermal resistance ( c/w). for on semiconductor components, the value for ja can be found on page 19 of the datasheet, under ?package thermal data.? note that this value is different for every package style and every manufacturer. for the ncp144x, ja varies between 10 ? 50 c/w, depending upon the size of the copper pad to which the ic is mounted. once the designer has calculated t j , the question of whether the ncp144x can be used in an application is settled. if t j exceeds 150 c, the absolute maximum allowable junction temperature, the ncp144x is not suitable for that application. if t j approaches 150 c, the designer should consider possible means of reducing the junction temperature. perhaps another converter topology could be selected to reduce the switch current. increasing the airflow across the surface of the chip might be considered to reduce t a . a copper ?landing pad? can be connected to ground ? designers are referred to on semiconductor applications note sr006 for more information on properly sizing a copper area. circuit layout guidelines in any switching power supply, circuit layout is very important for proper operation. rapidly switching currents combined with trace inductance generates voltage transitions that can cause problems. therefore the following guidelines should be followed in the layout. 1. in boost circuits, high ac current circulates within the loop composed of the diode, output capacitor, and on ? chip power transistor. the length of associated traces and leads should be kept as short as possible. in the flyback circuit, high ac current loops exist on both sides of the transformer. on the primary side, the loop consists of the input capacitor, transformer, and on ? chip power transistor, while the transformer, rectifier diodes, and output capacitors form another loop on the secondary side. just as in the boost circuit, all traces and leads containing large ac currents should be kept short. 2. separate the low current signal grounds from the power grounds. use single point grounding or ground plane construction for the best results. 3. locate the voltage feedback resistors as near the ic as possible to keep the sensitive feedback wiring short. connect feedback resistors to the low current analog ground.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 18 ordering information device operating temperature range package shipping ? ncp1442fr4 0 c < t a < 85 c 7 lead powerflex short ? leaded 2000 tape & reel ncp1442fr4g 7 lead powerflex short ? leaded (pb ? free) 2000 tape & reel ncp1442t 7 lead to ? 220 (straight lead) 50 units/rail ncp1443fr4 7 lead powerflex short ? leaded 2000 tape & reel ncp1443fr4g 7 lead powerflex short ? leaded (pb ? free) 2000 tape & reel ncp1443t 7 lead to ? 220 (straight lead) 50 units/rail ncp1444fr4 7 lead powerflex short ? leaded 2000 tape & reel ncp1444t 7 lead to ? 220 (straight lead) 50 units/rail ncp1445fr4 7 lead powerflex short ? leaded 2000 tape & reel ncp1445t 7 lead to ? 220 (straight lead) 50 units/rail ?for information on tape and reel specifications, including part orientation and tape sizes, please refer to our tape and reel packaging specifications brochure, brd8011/d.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 19 package dimensions powerflex 7 ? pin f suffix case 936j ? 01 issue o 3 6 notes: 1. dimensions and tolerancing per ansi y14.5m, 1982. 2. controlling dimension: inch. 3. dimensions a and b do not include mold flash or gate protrusions. mold flash and gate protrusions not to exceed 0.025 (0.635) max. a j l c d 7 pl g 0.076 (0.003) 7 pl b m n p dim a min max min max millimeters 0.350 0.360 8.89 9.14 inches b 0.350 0.360 8.89 9.14 c 0.070 0.080 1.78 2.03 d 0.026 0.030 0.66 0.76 e 0.005 0.015 0.13 0.38 f 0.031 0.041 0.79 1.04 g 0.050 bsc 1.270 bsc h 0.008 0.012 0.199 0.301 j 0.410 0.420 10.41 10.67 k 0.365 00.375 9.27 9.53 l 0.040 ref 1.02 ref m 0.367 9.16 9.31 n 0.310 0.320 7.87 8.13 p 0.394 0.400 10.00 10.16 r 0.002 ??? 0.05 ??? s 0.070 0.080 1.78 2.03 u 0.001 0.005 0.03 0.13 v w 0.296 ref 7.52 ref y 0.075 ref 1.91 ref aa 0.071 ref 1.81 ref ab 0.140 ref 3.56 ref ac 0.220 ref 5.58 ref ad 0.281 ref 7.14 ref ae 12 af 3 6 ae detail ag ? t ? seating plane s k y ad w aa ac ab thermal die pad af detail ag u f h e r (top offset) r 0.20 (0.008) r 0.25 (0.010) v 12 12 12 0.361 package thermal data parameter powerflex 7 ? pin unit r jc typical 1.0 ? 4.0 c/w r ja typical 10 ? 50* c/w *depending on thermal properties of substrate. r ja = r jc + r ca.
ncp1442, ncp1443, ncp1444, ncp1445 http://onsemi.com 20 package dimensions 7 lead to ? 220 t suffix case 821p ? 03 issue b notes: 1. dimensioning and tolerancing per ansi y14.5m, 1982. 2. controlling dimension: millimeter. 3. 821p?01 and ?02 obsolete. new standard is 821p?03. a f q d b c n u e j h 7 pl g 6 pl m k l dim min max min max inches millimeters a 9.91 10.54 0.390 0.415 b 8.23 9.40 0.324 0.370 c 4.19 4.83 0.165 0.190 d 0.66 0.81 0.026 0.032 e 0.89 1.40 0.035 0.055 f 7.62 typ 0.3 typ g 1.22 1.32 0.052 h 2.16 2.92 0.085 0.115 j 0.30 0.64 0.012 0.025 k 24.00 26.54 0.945 1.045 l 26.67 29.03 1.050 1.143 m 6.10 6.48 0.240 0.255 n 7 ??? 7 ??? q 3.53 3.96 0.139 0.156 u 4 6 4 6 0.048 on semiconductor and are registered trademarks of semiconductor components industries, llc (scillc). scillc reserves the right to mak e changes without further notice to any products herein. scillc makes no warranty, representation or guarantee regarding the suitability of its products for an y particular purpose, nor does scillc assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including wi thout limitation special, consequential or incidental damages. ?typical? parameters which may be provided in scillc data sheets and/or specifications can and do vary in different application s and actual performance may vary over time. all operating parameters, including ?typicals? must be validated for each customer application by customer?s technical experts. scillc does not convey any license under its patent rights nor the rights of others. scillc products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the scillc product could create a sit uation where personal injury or death may occur. should buyer purchase or use scillc products for any such unintended or unauthorized application, buyer shall indemnify and hold scillc and its of ficers, employees, subsidiaries, af filiates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, direct ly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that scillc was negligent regarding the design or manufacture of the part. scillc is an equal opportunity/affirmative action employer. this literature is subject to all applicable copyright laws and is not for resale in any manner. publication ordering information n. american technical support : 800 ? 282 ? 9855 toll free usa/canada europe, middle east and africa technical support: phone: 421 33 790 2910 japan customer focus center phone: 81 ? 3 ? 5773 ? 3850 ncp1442/d powerflex is a trademark of texas instruments incorporated. literature fulfillment : literature distribution center for on semiconductor p.o. box 5163, denver, colorado 80217 usa phone : 303 ? 675 ? 2175 or 800 ? 344 ? 3860 toll free usa/canada fax : 303 ? 675 ? 2176 or 800 ? 344 ? 3867 toll free usa/canada email : orderlit@onsemi.com on semiconductor website : www.onsemi.com order literature : http://www.onsemi.com/orderlit for additional information, please contact your local sales representative

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